Switching power supply apparatus

ABSTRACT

A switching power supply apparatus has a power circuit unit, the power circuit unit including a first switching element connected at a first end to a first end of a power supply, an inductor connected at a first end thereof to a second end of the first switching element, a second output terminal connected to a second end of the inductor, a capacitor connected between a first output terminal and the second output terminal, and a second switching element connected between a second end of the power supply and a second end of the first switching element. The switching power supply apparatus has an optimal response multi-mode digital current program mode control unit, the optimal response multi-mode digital current program mode control unit including an error signal generator which generates an error signal according to a potential difference between an output voltage and a preset voltage, and an inductor current detector which detects and amplifies an inductor current.

CROSS-REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority fromthe prior Japanese Patent Application No. 2010-34925, filed on Feb. 19,2010, the entire contents of which are incorporated herein by reference.

BACKGROUND

1. Field

Embodiments described herein relate generally to a switching powersupply apparatus.

2. Background Art

Conventionally, the switching power supply apparatus has been widelyutilized for a reason of its high power processing efficiency. Theswitching power supply apparatus is a power supply apparatus including aswitching element, a smoothing circuit formed of an inductor and acapacitor, and a feedback control circuit for keeping an output powersupply voltage constant. As for its operation, a smoothing circuitsmoothes a rectangular wave obtained by connecting a high potentialterminal and a low potential terminal of a supply source alternately bymeans of the switching element, and an output power supply having avoltage, which is different from that of the supply source is obtained.The switching element assumes extremely low resistance when it is in theon-state, whereas the switching element assumes extremely highresistance when it is in the off-state. In both cases, power dissipationin the switching element is very low. In addition, ideally, the inductorand capacitor do not dissipate power in their charging and dischargingoperation. Therefore, the switching power supply apparatus has a featurethat the power dissipation is low.

As for the feedback control circuit in the switching power supplyapparatus, either a voltage feedback control mode or a current programmode is mainly used (see, for example, U.S. Patent Application No.2009/0267582).

The voltage feedback control mode is a voltage feedback pulse-widthmodulation mode in which a voltage of an output power supply is measuredand a time ratio between the on-state and off-state of the switchingelement is changed according to a difference between the measuredvoltage and a reference potential. As a matter of fact, however, asignificant transfer delay is caused in the smoothing circuit formed ofthe inductor and capacitor in the operation of the voltage feedbackcontrol mode. Therefore, it is difficult to stabilize the controloperation.

On the other hand, in the current program mode controls, the peak ofinductor current in which a current flowing through an inductor ismeasured and the switching element is kept in the on-state until theinductor current reaches a certain target value and then the switchingelement is brought into the off-state. The target value of the inductorcurrent used here is calculated from a difference between a voltage ofthe output power supply and a reference potential. Its feedback controlbecomes a double-loop structure and is complicated. However, the delayin the smoothing circuit, which poses a problem in voltage-modecontrollers, is significantly reduced in the current program mode.Therefore, it is comparatively easy to stabilize its control operation.

As an important characteristic in the switching power supply apparatus,there is a transitional response characteristic (also known as transientresponse) which appears in the case where the load current has changedremarkably and rapidly. If the transitional response characteristic isslow, then a phenomenon that the output voltage becomes low in a momentoccurs, for example, when a state in which the load current is small hasabruptly changed to a state in which the load current is large.Conversely, when a state in which the load current is large has abruptlychanged to a state in which the load current is small, the outputvoltage rises in a moment.

In the voltage feedback control mode, nonlinear control is frequentlyused. In the nonlinear control, a difference between the output voltageand the preset voltage caused by a change of the load current isdetected and the on-time of the switching element is prolonged extremelyto cancel the difference. In the current program mode, however, it isdifficult to adopt the nonlinear control, therefore the transitionalresponse characteristic obtained when the load current has varied oftenposes a problem especially in the switching power supply apparatus usingthe current program mode.

In this way, the switching power supply apparatus using the conventionalvoltage feedback control mode has a problem that it is difficult toensure the stability. Furthermore, in the switching power supplyapparatus using the conventional voltage feedback control mode, paralleloperation is challenging to achieve and it is difficult to shrink thesize of the apparatus and reduce the cost of the apparatus. In addition,problems with non-equal current sharing between phases, causingpotential component failures is quite common.

On the other hand, in the switching power supply apparatus using theconventional current program mode, which is an improved type for theswitching power supply apparatus using the conventional voltage feedbackcontrol mode, there is a problem that it is difficult to improve thetransitional response characteristic.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the switching power supply apparatus;

FIG. 2 is a waveform diagram of optimal transient response operation inthe actual switching power supply apparatus according to the firstembodiment;

FIG. 3 is a zoomed in version of the waveform diagram of FIG. 2 with thetime axis expanded;

FIG. 4 is a waveform diagram of a pulse frequency control operation in alow load state in the actual switching power supply apparatus accordingto the first embodiment;

FIG. 5 is a block diagram showing the optimal response multi-modedigital current program mode control unit, which is the principal partof the switching power supply apparatus;

FIG. 6 is a waveform diagram for explaining operation of the switchingpower supply apparatus;

FIG. 7 is an operation waveform diagram of an actual switching powersupply apparatus;

FIG. 8 is a block diagram showing a switching power supply apparatusaccording to the third embodiment of the present invention;

FIG. 9 is a diagram for explaining a problem of voltage control modevoltage step-down power supplies connected in parallel; and

FIG. 10 is a concept diagram of current program mode voltage step-downpower supplies connected in parallel according to the third embodimentof the present invention.

DETAILED DESCRIPTION

A switching power supply apparatus according to an embodiment, includesa power circuit unit, the power circuit unit including a first switchingelement connected at a first end to a first end of a power supply, aninductor connected at a first end thereof to a second end of the firstswitching element, a second output terminal connected to a second end ofthe inductor, a capacitor connected between a first output terminal andthe second output terminal, and a second switching element connectedbetween a second end of the power supply and a second end of the firstswitching element.

The switching power supply apparatus further includes: an optimalresponse multi-mode digital current program mode control unit, theoptimal response multi-mode digital current program mode control unitincluding an error signal generator which generates an error signalaccording to a potential difference between an output voltage and apreset voltage (also known as reference), and an inductor currentdetector which detects and amplifies an inductor current.

The optimal response multi-mode digital current program mode controlunit includes: a mode detector which detects an abrupt change of theoutput voltage and switches an operation mode; a peak/valley detector &optimal current difference calculator which detects an extreme value ofthe output voltage and calculates a charge quantity lost by thecapacitor; and an inductor current detection storage circuit whichdetects and stores an inductor current value obtained when an extreme(for example, peak or valley) value of the output voltage is detected.

The optimal response multi-mode digital current program mode controlunit exercises on-off control of the first switching element or thesecond switching element according to control of the mode detector, thepeak/valley detector & optimal current difference calculator, and theinductor current detection storage circuit.

Hereafter, embodiments of the present invention will be described withreference to the drawings.

First Embodiment

First, a switching power supply apparatus (DC-DC converter) according toa first embodiment of the present invention will now be described withreference to the drawings. FIG. 1 is a block diagram showing theswitching power supply apparatus. FIG. 2 is a waveform diagram ofoptimal transient response operation in the actual switching powersupply apparatus according to the first embodiment. FIG. 3 is a waveformdiagram with a time axis in FIG. 2 expanded. FIG. 4 is a waveformdiagram of a pulse frequency control operation in a low load state inthe actual switching power supply apparatus according to the firstembodiment.

The switching power supply apparatus according to the present embodimentis a DC voltage step-down power supply which operates by using anoptimal response multi-mode digital current program mode controllerhaving a self-aligning function.

As shown in FIG. 1, the switching power supply apparatus includes anoptimal response multi-mode digital current program mode control unitand a power circuit unit, which operates in response to a signalsupplied from the optimal response multi-mode digital current programmode control unit. The switching power supply control apparatus forcontrolling the power circuit unit includes the optimal responsemulti-mode digital current program mode control unit. In thisembodiment, for example, the switching power supply control apparatus isequal to the optimal response multi-mode digital current program modecontrol unit.

The power circuit unit includes a first switching element (MOStransistor) Q1, a second switching element (MOS transistor) Q2, aninductor L and a capacitor C. A high potential terminal + of an externalinput power supply v_(g)(t) is connected to a first terminal of thefirst switching element Q1, and a low potential terminal − of theexternal input power supply v_(g)(t) is connected to a first terminal ofthe second switching element Q2. A second terminal of the firstswitching element Q1 and a second terminal of the second switchingelement Q2 are connected to each other, and connected in common to afirst terminal of the inductor L. A second terminal of the inductor Lbecomes a high potential output terminal + of the DC voltage step-downpower supply. The low potential terminal − of the external input powersupply v_(g)(t) becomes a low potential output terminal − of the DCvoltage step-down power supply. The capacitor C is connected between thehigh potential output terminal + and the low potential output terminal −of the DC voltage stepdown power supply. A load is also connectedbetween the high potential output terminal + and the low potentialoutput terminal − of the DC voltage step-down power supply.

Switching signals, which are output from the optimal response multi-modedigital current program mode control unit are coupled to gate terminalsof the first switching element Q1 and the second switching element Q2via digital buffer circuits, respectively. Furthermore, in thisembodiment, a sum of a current which flows through the first switchingelement Q1 and a current which flows through the second switchingelement Q2 and an output voltage of the DC voltage stepdown power supplyare always monitored by the optimal response multi-mode digital currentprogram mode control unit.

When forming a current detection circuit, there is a problem to besolved. Conventionally, means for detecting a current which flowsthrough the inductor has been adopted. Since a delay is caused in thedetected current value because of influence of parasitic capacitance ofthe inductor, however, it is not optimal to use the means for detectingthe current which flows through the inductor in the peak current controlmode. In the peak current control mode, therefore, it is demanded todetect the current which flows through the first switching element Q1.However, the current becomes discontinuous, and an amplifier is requiredto have a bandwidth which is so high that implementation is difficult,to detect such a current. In the present embodiment, this problem issolved by using means which adds the current flowing through the firstswitching element Q1 and the current flowing through the secondswitching element Q2 and amplifies a result of the addition. In otherwords, the discontinuous inductor current is reconstructed to have acontinuous state and the demand for the bandwidth upon an inductorcurrent amplifier 7 in the current detection circuit is relaxed.

The optimal response multi-mode digital current program mode controlunit is a principal part in the present embodiment, and it includes apeak/valley detector & optimal Δi_(c)[n] calculator 1, a mode detector2, an error multiplier 3, a first mode switch 4, a successiveapproximation DA converter 5, an inductor current adder 6, the inductorcurrent amplifier 7, an inductor current sample and hold circuit 8, asecond mode switch 9, an optimal current value retaining capacitor 10, adifferential amplifier 11, an SR latch 12, a clock generator 13, and adead time & V_(drive) generator 14.

Incidentally, the peak/valley detector & optimal Δi_(c)[n] calculator 1,the mode detector 2, the error multiplier 3, and the first mode switch 4configure an output potential monitor. Furthermore, the inductor currentadder 6, the inductor current amplifier 7, the inductor current sampleand hold circuit 8, the second mode switch 9, and the optimal currentvalue retaining capacitor 10 form an inductor current detection storagecircuit.

In this embodiment, the high potential output terminal + of the DCvoltage stepdown power supply becomes an input of an asynchronous flashanalog-digital converter 1-1 included in the peak/valley detector &optimal Δi_(c)[n] calculator 1.

The asynchronous flash analog-digital converter 1-1 compares a potentialat the high potential output terminal + of the DC voltage stepdown powersupply with a given reference voltage V_(ref), and outputs a deviationquantity from the preset potential, i.e., an error signal e[n]. A ringoscillators & digital logic circuit 1-2, which is a principal part ofthe peak/valley detector & optimal Δi_(c)[n] calculator 1, generates apeak/valley detection signal s(t) and an optimal peak current differencesignal Δi_(c)[n] based on the error signal e[n]. The error signal e[n]is supplied to the mode detector 2 and the error multiplier 3.

The mode detector 2 outputs a mode changeover signal m(t) based onmagnitude of the error signal e[n]. The first mode switch 4 (sw1)selects either an output Ke[n] of the error multiplier 3 or an output ofthe peak/valley detector & optimal Δi_(c)[n] calculator 1 according toan order of the mode changeover signal m(t), and calculates the latestoptimal peak current difference signal Δi_(c)[n].

The optimal peak current difference signal Δi_(c)[n] is input to anoutput adder 5-1 included in the successive approximation DA converter5.

The output adder 5-1 adds together the optimal peak current differencesignal Δi_(c)[n] and an optimal peak current signal i_(c)[n−1] of thelast time, which is output from an output analog-digital converter 5-2,and calculates the present optimal peak current signal i_(c)[n]. Thecalculated latest optimal peak current signal i_(c)[n] is converted toan analog signal voltage signal R_(c)i_(c)(t) by an outputdigital-analog converter 5-3, and the analog signal voltage signalR_(c)i_(c)(t) is output from the successive approximation DA converter5.

On the other hand, detected signals respectively corresponding to thecurrent which flows through the first switching element Q1 and thecurrent which flows through the second switching element Q2 are addedtogether by the inductor current adder 6 to produce an analog signali_(L)(t) which represents an inductor current. The inductor currentsignal i_(L)(t) is amplified to R_(s) times by the inductor currentamplifier 7 to produce an inductor current signal R_(s)i_(L)(t). Theinductor current signal R_(s)i_(L)(t) is taken into and retained by theinductor current sample and hold circuit 8 according to an order of thepeak/valley detection signal s(t), which is output from the peak/valleydetector & optimal Δi_(c)[n] calculator 1.

The second mode switch 9 selects and outputs either the inductor currentsignal R_(s)i_(L)(t) which is output from the inductor current sampleand hold circuit 8 or the output signal R_(c)i_(c)(t) of the successiveapproximation DA converter 5 according to the order of the peak/valleydetection signal s(t). The selected output signal is stored across theoptimal current value retaining capacitor 10 as the latest optimal peakcurrent value R_(c)i_(c)(t). The optimal peak current valueR_(c)i_(c)(t) stored across the optimal current value retainingcapacitor 10 is supplied to the output analog-digital converter 5-2.

The output analog-digital converter 5-2 converts the analog signalR_(c)i_(c)(t), which indicates the optimal peak current value, to adigital signal. The digital signal of the optimal peak current value isreferred to in the next cycle as the last optimal peak current valuei_(c)[n−1].

The measured inductor current signal R_(s)i_(L)(t) is input to anon-inverting input terminal + of the differential amplifier 11, whereasthe optimal peak current value R_(c)i_(c)(t) is input to an invertinginput terminal − of the differential amplifier 11. When the inductorcurrent signal R_(s)i_(L)(t) has reached the optimal peak current valueR_(c)i_(c)(t), the differential amplifier 11 outputs a reset signal R.

The reset signal R is coupled to a reset terminal R of the SR latch 12.A clock signal generated by the clock generator 13 is coupled to a setterminal of the SR latch 12. The SR latch 12 receives the set signal Sand the reset signal R, and outputs a control signal c(t). The dead time& V_(drive) generator 14 is supplied with the control signal c(t), andoutputs drive signals which control on/off of the first switchingelement Q1 and the second switching element Q2. At this time, the deadtime & V_(drive) generator 14 conducts optimization of a time periodover which both the first switching element Q1 and the second switchingelement Q2 are turned off, i.e., adjustment of the dead time based onthe optimal peak current signal i_(c)[n] in the successive approximationDA converter 5.

The dead time & V_(drive) generator 14 converts a voltage (notillustrated) of a power supply supplied to the optimal responsemulti-mode digital current program mode control unit to a voltageV_(g)(t) of a power supply supplied to the power circuit unit, and thenoutputs the drive signals.

The switching power supply apparatus having a configuration describedheretofore conducts two different operations according to the magnitudeof the load current. When the load current is great in the steady state,the switching power supply apparatus conducts a current program typepulse-width modulation operation. When the load current is small in thesteady state, the switching power supply apparatus conducts apulse-frequency modulation operation. When the load current is rapidlychanging in the transitional state, the switching power supply apparatusconducts an optimal transitional response operation. Hereafter,respective operations will be described.

First, the current program type pulse width modulation operationconducted when the load current is larger than a preset value, in thesteady state will now be described.

The asynchronous flash analog-digital converter 1-1 compares the outputvoltage with the reference voltage V_(ref), and generates the digitalerror signal e[n]. The digital error signal e[n] is supplied to the modedetector 2. When the absolute value of the digital error signal e[n] issmall, for example, when |e[n]|≦1, the first mode switch 4 (sw1) and thesecond mode switch 9 (sw2) are set to a position 1 side. At this time,the load current is determined to be in the steady state, and thecurrent program type pulse width modulation operation, which is thefirst operation state, or the pulse frequency modulation operation,which is the second operation state, is conducted. In both operationstates, the digital error signal e[n] is supplied to the errormultiplier 3. The error multiplier 3 forms an error compensationcalculator in conjunction with an analog-digital converter whichincludes the successive approximation DA converter 5. The errorcompensation calculator calculates the peak current digital signali_(c)[n] according to Equation (1).

i _(c) [n]=i _(c) [n−1]+Ke[n]  (1)

The calculated peak current digital signal i_(c)[n] is converted to thepeak current analog signal R_(c)i_(c)(t). The peak current analog signalR_(c)i_(c)(t) is compared with the inductor current analog signalR_(s)i_(L)(t). When the inductor current analog signal R_(s)i_(L)(t)output from the inductor current detection circuit has become greaterthan the peak current analog signal R_(c)i_(c)(t), the SR latch 12 isreset. The SR latch 12 receives the clock signal from the clockgenerator 13. The SR latch 12 is set at the beginning of each switchingcycle to generate a pulse-width-modulated control signal c(t).

The pulse frequency modulation operation which is the second operationstate will now be described.

The pulse frequency modulation operation is nearly the same as the pulsewidth modulation operation which is the first operation state. If thecalculated peak current signal i_(c)[n] has become less than apredetermined PFM current threshold i_(pfm)[n], then the clock generator13 is brought into the stop state and the output of the successiveapproximation DA converter 5 is set to a fixed value which is lower thanthe ripple current of the inductor. Furthermore, upon receiving anoutput signal of the asynchronous flash analog-digital converter 1-1,which is not illustrated, the SR latch 12 sets its output signal c(t).

The control form of the output voltage is an improved hysteresis mode ofsome kind. When a deviation between the output voltage and a presetvalue, i.e., the error signal e[n] has exceeded 1, the asynchronousflash analog-digital converter 1-1 activates the SR latch 12. Thereupon,the first switching element Q1 is brought into the conduction state, andthe inductor current increases. As a result, the output voltage rises.When the inductor current has reached a fixed value which is set in thesuccessive approximation DA converter 5, the SR switch 12 is inactivatedand the first switching element Q1 is brought into the cutoff state. Theoutput capacitance is discharged by the load current. When the errorsignal e[n] has exceeded 1 again, the next switching period is started.

Finally, the optimal load response operation which is the thirdoperation state will be described.

The load variation is detected by the mode detector 2.

Specifically, at the instant when the absolute value of the differencebetween the output voltage and the preset voltage exceeds unity, i.e.,the relation |e[n]|≧2 is satisfied, the optimal load response operationis conducted. For example, at the instant when the transition from alight load state to a heavy load state is detected, the first switchingelement Q1 is brought into the conduction state. Conversely, if the loadtransition is occurring in the opposite direction, the first switchingelement Q1 is brought into the cutoff state. At that time, the firstmode switch 4 (sw1) is set to a second state (a position 2 side) by themode changeover signal m(t), and the peak/valley detector & optimalΔi_(c)[n] calculator 1 is activated.

The peak/valley detector & optimal Δi_(c)[n] calculator 1 calculates theoptimal peak value of the inductor current according to a calculationtechnique based on the capacitor charge balance method. According to thecalculation technique, the lost charge quantity can be restored mostquickly until the optimal value in the steady state after thetransitional response operation is reached. Furthermore, in the process,a rebound is not caused in the output voltage. Conventionally, however,there is a problem that the calculation precision of the calculationtechnique is influenced intensely by variation of the inductor includedin the power circuit unit, variation of the capacitor, and a loss causedby parasitic elements. In the present embodiment, the change quantity ofthe inductor current with time during the transitional time period iscorrected by using the inductor current sample and hold circuit 8.

Incidentally, operation of the capacitor charge balance method and thetechnique for correcting the influence of the variation of the inductor,the variation of the capacitor, and the loss caused by parasiticelements will be described below in detail with reference to a secondembodiment.

Finally, effects of the present embodiment will be described withreference to an operation waveform diagram of an actual switching powersupply apparatus configured based on the present embodiment.

Operation waveform diagrams in FIGS. 2 to 4 show measurement results ofvoltage stepdown operation for obtaining an output power supply of 1.5 Vfrom an external power supply of 7.5 V in the present embodiment. Itsoperation frequency is 500 kHz.

FIG. 2 and FIG. 3 which is an expanded diagram of FIG. 2 show theoptimal transitional response operation obtained when the load currenthas changed from 1.5 A to 4.5 A. FIG. 2 shows the output power supplyvoltage on the scale of 200 mV/division, the inductor current on thescale of 2 A/division, the current control reference value on the scaleof 5 A/division, and the load current controlling a command signal onthe scale of 10 V/division, in order from the top toward the bottom. Thetime axis of all of them has the scale of 20 μs/division. FIG. 3 is adiagram of the same waveforms obtained by expanding the time axis. FIG.3 shows the output power supply voltage on the scale of 100 mV/division,the inductor current on the scale of 2 A/division, the current controlreference value on the scale of 5 A/division, and the load currentcontrolling a command signal on the scale of 10 V/division. The timeaxis of all of them has the scale of 5 μs/division. As appreciated fromthe waveform diagrams, the voltage of the output power supply returns tothe steady state after the on/off operation is conducted only once, i.e.in the optimal time. According to the control mode in the presentembodiment, it is possible to provide a voltage step-down power supplycapable of having a transitional response that is very fast as comparedwith the conventional control mode, in this way.

FIG. 4 shows the steady state operation of the pulse frequencymodulation in a control circuit configured based on a proposal of thepresent embodiment. FIG. 4 shows the output power supply voltage on thescale of 1 V/division, the inductor current on the scale of 2A/division, and the switch element gate drive signal on the scale of 5V/division in order from the top toward the bottom. The time axis of allof them has the scale of 5 μs/division. In this steady state operationof the pulse frequency modulation, the load current is 150 mA. When thefirst switch element Q1 is in the conduction state, the inductor currentincreases rapidly. Conversely, when the second switch element Q2 is inthe conduction state, the inductor current decreases rapidly. For sometime, both the first switch element Q1 and the second switch element Q2are brought into the cutoff state and the inductor current becomesnearly zero. In other words, a discontinuous conduction mode (DCM) isbrought about, and it is a mode, which is effective to reduce the powerdissipation of the power supply circuit under the low load operationcondition. The matter that poses a problem at this time is the stabilityof the voltage of the output power supply. It is appreciated that theoutput voltage is stable at 1.5 V as shown in FIG. 4. If the controlcircuit configured based on the present embodiment is used, then in thisway it is possible to implement the pulse frequency modulation steadystate operation with a smaller circuit scale as compared with theconventional mode and provide a voltage stepdown power supplydissipating low power at a lower cost.

In implementing the three functions, i.e., the pulse width modulationoperation, the pulse frequency modulation operation and the optimaltransitional response operation, virtually no additional elements formulti-mode operation are needed unlike the case of the conventionalvoltage controlled switching power supply apparatus. Therefore, a newincrease of the manufacturing cost required for the control circuit canbe minimized.

The conventional current controlled switching power supply apparatus hasa problem that it is difficult to raise the speed of control forsuppressing the voltage variation of the output power supply when theload has changed abruptly. This is because introduction of nonlinearcontrol frequently used in the voltage controlled switching power supplyapparatus is difficult in the current controlled switching power supplyapparatus. Conventionally, therefore, means which switches the controlmode to the voltage control mode when a variation is caused in outputvoltage by a sudden change of the load and returns the control mode tothe current control mode again when the load has become stable is used.In this case, however, discontinuity occurs when the control mode isswitched and the control becomes unstable. Furthermore, recently thereis also a proposal of a current control mode utilizing the capacitorcharge balance method in the same way as the present embodiment.However, the influence of the variation of the inductor, the variationof the capacitor, and the loss caused by parasitic elements appearsremarkably, and consequently the voltage of the output power supply doesnot become stable.

According to the present embodiment, stable operation is demonstrated inpeak current control, in the pulse width modulation operation, as wellas in the pulse frequency modulation operation and the optimaltransitional response operation. In addition, seamless transitionbetween the control modes is achieved. The inductor current is detectedand retained in each of the operation states, and it is used as anoptimal peak current value in the next operation. Therefore, the controldoes not become discontinuous at the time of switching of the operationstate. Furthermore, a function of correcting the influence of thevariation of the inductor, the variation of the capacitor, and the losscaused by parasitic elements in the power circuit unit is provided inthe operation for detecting and retaining the inductor current.According to the present embodiment, therefore, it is possible toprovide a switching power supply apparatus in which the voltage isstable against an abrupt load variation.

Second Embodiment

A switching power supply apparatus (DC-DC converter) according to asecond embodiment of the present invention will now be described withreference to the drawings.

FIG. 5 is a block diagram showing the optimal response multi-modedigital current program mode control unit, which is the principal partof the switching power supply apparatus. FIG. 6 is a waveform diagramfor explaining operation of the switching power supply apparatus. FIG. 7is an operation waveform diagram of an actual switching power supplyapparatus. The present embodiment provides an optimal responsemulti-mode digital current program mode controller having a self-alignedfunction, which is used in the DC power supply.

The optimal response multi-mode digital current program mode controllershown in FIG. 5 can form a switching power supply apparatus by combiningit with the power circuit unit in the same way as the first embodiment.Its configuration is nearly the same as that in the first embodiment.Hereafter, therefore, the same components as those in the firstembodiment are denoted by like characters and description thereof willnot be repeated. Only different parts will be described.

The optimal response multi-mode digital current program mode controllershown in FIG. 5 is formed by adding a programmable current protection, alimiter 15, and a reset signal compounder 16 to the optimal responsemulti-mode digital current program mode controller in the switchingpower supply apparatus shown in FIG. 1. They are elements provided toprevent the inductor current from being saturated and elements frombeing destroyed. In the conventional optimal transitional responsecontrol mode based on the capacitor charge balance method, the on-timeand off-time of the switch are calculated. Therefore, there is a problemthat the on-time and the off-time are influenced by the inductormanufacturing dispersion, current saturation and parasitic resistanceand its correction is very difficult. In addition, the inductorovercurrent protection and optimal transitional response control arecontradicting functions, and it has been considered to reconcile them.According to the current program mode control circuit based on thepresent embodiment, the function concerning such current control can beimplemented by adding a comparatively small number of circuit parts.

Hereafter, operation of the capacitor charge balance method implementingthe optimal transitional response, which is a feature of the presentembodiment and the technique for correcting the influence of thevariation of the inductor, the variation of the capacitor, and the losscaused by parasitic elements will be described in detail with referenceto an operation waveform diagram shown in FIG. 6.

FIG. 6 shows waveforms of the output voltage v(t) and inductor currenti_(L)(t) at the time of optimal transitional response operation when atransition from a small load current state to a large load current stateis made. Besides the inductor current waveform i_(L)(t), a referencevalue of the current control, i.e., a threshold is shown. Incidentally,it is a matter of course that a similar load response operation is alsopossible in the case where a transition from the large load currentstate to the small load current state is made.

At the instant when a load variation is detected, the first switchingelement Q1 is brought into the conduction state and an optimaltransitional response control circuit is activated. The optimaltransitional response control circuit sets a maximum transitionalresponse current, which is denoted by i_(pk) in FIG. 6 and sets a steadyvalue after the transition, denoted by i_(ctrl) _(—) _(new) in FIG. 6.According to a feature of the capacitor charge balance method providedby the present embodiment, only the maximum transitional responsecurrent i_(pk) is calculated and the steady value i_(ctrl) _(—) _(new)after the transition is measured and retained in the inductor currentsample and hold circuit 8 shown in FIGS. 1 and 5 during the transitionoperation. At the instant when a minimum point (valley point), whichassumes an extreme value (a minimum value) of the output voltage v(t) isdetected as shown in FIG. 6. A sampling pulse signal having a shortpulse width denoted by s(t) in FIGS. 1 and 5 is generated and theinductor current at timing of generation of the minimum point iscaptured by a capacitor C_(DAC) 10 included in a filter. As shown in thewaveform diagram of FIG. 6, the inductor current becomes the steadyvalue i_(ctrl) _(—) _(new) after the transition. At the same time, themaximum transitional response current i_(pk) is calculated from anoptimal current difference Δi[n] according to Equation (2) as shown inthe waveform diagram of FIG. 6.

i _(pk) [n]=i _(ctrl) _(—) _(new) [n]+Δi[n]  (2)

An output digital-analog converter 5-3 (Σ-ΔDAC) operates as afirst-order sigma-delta modulator and a kind of RC filter circuit. Inthe steady state, an output analog-digital converter 5-2 (flash ADC)conducts sampling operation at a frequency, which is lower than theswitching frequency in order to minimize the controller powerdissipation, and only R1 in a first filter circuit resistor 5-4 is used.On the other hand, during a time period when the calculation of Equation(2) is executed, i.e. during load transients, operation frequencies ofthe output analog-digital converter 5-2 and the output digital-analogconverter 5-3 are raised and the first filter circuit resistor 5-4 (R1)and the second filter circuit resistor 5-5 (R2) are connected inparallel. As a result, the cutoff frequency of the RC filter is raised.At the same time, the speed of the digital-analog conversion is raisedand the maximum transitional response current ipk can be found beforethe inductor current iL actually assumes its value.

In the capacitor charge balance method according to the conventionalart, instability caused in a shift from the optimal transitionalresponse operation to the steady operation poses a great problem. Thisis caused by the fact that a technique of calculating the conductiontime ton and the cutoff time toff of the first switching element asshown in FIG. 6 is used in the conventional capacitor charge balancemethod. The conduction time ton and cutoff time toff of the actualswitching element deviate significantly from calculated values becausethey are influenced intensely by variations of the inductor andcapacitor and the loss caused by the parasitic elements. On the otherhand, according to the present embodiment, the inductor current steadyvalue ictrl_new after the transition is acquired by a detector circuit.As a result, the problem of stability caused in the shift from theoptimal transitional response operation to the steady operation iscompletely dissolved.

The reason why the present embodiment is stable under the influence ofthe variations of the inductor and the capacitor and the loss caused bythe parasitic elements will now be described.

A capacitance charge balance algorithm is used to calculate the currentdifference Δi. Charge Q lost from the capacitor C, which has caused thevoltage variation, must be replenished extra by the inductor current. Inother words, Equation (3) holds true.

$\begin{matrix}{Q = {{C\; \Delta \; v} = {\frac{1}{2}\Delta \; {i( {t_{on} + t_{off}} )}}}} & (3)\end{matrix}$

Here, relations between the optimal on-time ton and the optimal off-timetoff of the switch and the current difference Δi are given by Equation(4).

Δi=a_(r)t_(on) and Δi=a_(f)t_(off)  (4)

Here, ar and af are rising and falling slopes of the inductor current,respectively. As for the current difference Δi, Equation (5) is derivedby combining Equations (3) and (4). In Equation (5), Δv represents apotential difference between the steady value and the minimum value ofthe output voltage.

$\begin{matrix}{{\Delta \; i} = \sqrt{\frac{2C\; \Delta \; {v \cdot a_{r}}a_{f}}{a_{r} + a_{f}}}} & (5)\end{matrix}$

In an ideal lossless voltage stepdown power supply, the rising andfalling slopes of the current become as shown in the operation waveformdiagram in FIG. 6. Their relations are represented by Equation (6). InEquation (6), Vg represents a voltage of an external input power supply,and V represents the output voltage.

$\begin{matrix}{a_{r} = {{\frac{V_{g} - V}{L}\mspace{14mu} {and}\mspace{14mu} a_{f}} = \frac{V}{L}}} & (6)\end{matrix}$

Substituting Equation (6) into Equation (5) and rewriting, Equation (7)representing a current difference Δi in the ideal voltage stepdown powersupply is obtained. In Equation (7), D′ represents 1−D (where Drepresents a duty ratio).

$\begin{matrix}{{\Delta \; i} = \sqrt{\frac{2{CD}^{\prime}\Delta \; v}{L}}} & (7)\end{matrix}$

However, Equation (7) holds true only when the voltage stepdown powersupply can be regarded as a substantially ideal voltage stepdown powersupply as in the case where the voltage of the output power supply iscomparatively high and its load current is small. Otherwise, the resultcalculated supposing an ideal power supply deviates from the actualvalue largely, resulting in a response that cannot be said to beoptimal.

On the other hand, in the control configuration according to the presentembodiment, the inductor current settles down to a value in the optimalsteady state without fail in the final stage of the on/off operationeven if the voltage stepdown power supply is not optimal. This isbecause the current value in the latest steady state is found not from acalculation equation but by measuring, i.e. sampling, the actualinductor current at the point when a peak or the lowest point of thevoltage is reached as described heretofore.

One of features of the present embodiment is that it has a self-aligningfunction of the circuit constants L and C. Owing to the function,characteristic degradation of the voltage stepdown circuit caused by therising slope of the inductor current is compensated at the time of thetransition from the low load state to the high load state. In otherwords, a principal action of the self-aligning function is to find therising slope and the falling slope of the inductor current accurately.

In order to find the rising slope of the inductor current, the firston-time t_(on) of the transistor is measured by a ring oscillators &digital logic circuit 1-2, which is a principal part of the presentembodiment in the first transitional response. As for its method, theinductor current difference Δi is set equal to an expected highest valueΔi_(init) and time required for the inductor current to reach the valueis measured. The rising slope a_(r) of the inductor current can becalculated from the obtained result according to Equation (8).

$\begin{matrix}{a_{r} = \frac{\Delta \; i_{init}}{t_{on}}} & (8)\end{matrix}$

In the same way, the off-time of the switch or time required for thecontrol circuit to return from the highest current value to the lateststeady state current value is measured, and the falling slope of theinductor current is found according to Equation (9).

$\begin{matrix}{a_{f} = \frac{\Delta \; i_{init}}{t_{off}}} & (9)\end{matrix}$

Owing to such manipulation, all parameters included in Equation (5),except the capacitor value, can be calculated. Finally, in order to findthe capacitor value C, the difference between the output voltage and thepreset value is measured at the time point of end of the first optimaltransitional response operation. If the value is not zero, the capacitorvalue is corrected accordingly. In other words, owing to this process,all parameters in Equation (5) can be completely determined after thetransitional response operation is conducted once. The parameters in thecontrol circuit are adjusted into a state in which an optimal responsecan be obtained in both the current and the voltage, before the nextload variation occurs.

Finally, effects of the present embodiment will now be described withreference to an operation waveform diagram of the actual switching powersupply apparatus configured based on the present embodiment.

FIG. 7 is a diagram showing behavior of a current limiting function,which is a feature of the present embodiment. The operation waveformdiagram in FIG. 7 shows measurement results of a voltage stepdownoperation for obtaining an output power supply of 1.5 V from an externalpower supply of 7.5 V in the present embodiment. Its operation frequencyis 500 kHz. FIG. 7 shows a voltage waveform of the output power supplyon the scale of 200 mV/division, the inductor current on the scale of 2A/division, the current control reference value on the scale of 5A/division, and the load current controlling command signal on the scaleof 10 V/division, in order from the top toward the bottom. The time axisof all of them has the scale of 20 μs/division. Incidentally, thiscurrent limiting function is implemented by the programmable currentprotection and limiter 15, as well as the reset signal compounder 16shown in FIG. 15. Its role is to prevent the inductor current from beingsaturated. As for its operation, the first switch element Q1 is broughtinto the cutoff state for some time when a preset current limit isreached, and then the first switch element Q1 is brought into theconduction state again. As a result, control is exercised to prevent theinductor current from exceeding a maximum allowable current, and thecontrol operation is repeated until the charge balance is achieved.According to the control mode based on the present embodiment, it ispossible in this way to provide a voltage stepdown power supply capableof having a fast transitional response while providing a maximumallowable current limit.

Third Embodiment

A switching power supply apparatus (DC-DC converter) according to athird embodiment of the present invention will now be described withreference to the drawings. FIG. 8 is a block diagram showing a switchingpower supply apparatus according to the third embodiment of the presentinvention. FIG. 9 is a diagram for explaining a problem of voltagecontrol mode voltage stepdown power supplies connected in parallel. FIG.10 is a concept diagram of current program mode voltage stepdown powersupplies connected in parallel according to the third embodiment of thepresent invention.

The present embodiment is a DC high voltage power supply obtained byconnecting a plurality of DC voltage stepdown power supplies operated byan optimal response multi-mode digital current program mode control unithaving a self-aligning function, in parallel. It is attempted to obtaina large supply current by connecting plurality of DC voltage stepdownpower supplies in parallel in this way and using small-sized elements,especially small inductors.

The switching power supply apparatus shown in FIG. 8 has aconfiguration, which is nearly the same as that of the first embodiment.The same components as those in the first embodiment are denoted by likecharacters, and description of them will not be repeated and onlydifferent parts will be described. In FIG. 8, a plurality of systems ofa first switching element Q1, a second switching element Q2 and aninductor L in the power circuit unit, which are elements surrounded by adashed line, are prepared, and a common output power supply v(t) isgenerated from a common external input power supply v_(g)(t). In thesame way, a plurality of systems of a successive approximation DAconverter 5, an inductor current adder 6, the inductor current amplifier7, an inductor current sample and hold circuit 8, a second mode switch9, an optimal current value retaining capacitor 10, a differentialamplifier 11, an SR latch 12, and a dead time & V_(drive) generator 14included in an optimal response multi-mode digital current program modecontrol unit, which are elements surrounded by a dashed line in FIG. 8are prepared. The inductor current in each of the power circuit units isobserved, and the first switching element Q1 and the second switchingelement Q2 are driven. Other elements, i.e., a peak/valley detector &optimal Δi_(c)[n] calculator 1, a mode detector 2, an error multiplier3, a first mode switch 4, and a clock generator 13 are prepared by ones,and are used in common.

In the conventional switching power supply apparatus of the voltagefeedback control mode, it is basically difficult to implement amulti-phase operation in which a plurality of power supply circuits areconnected in parallel and caused to operate. This is because when theswitching power supply apparatuses of the voltage feedback control modeare connected in parallel as shown in FIG. 9 a phenomenon that currentconcentrate into a switching power supply apparatus of one system isbrought about by differences among components, for example, differencesamong inductors L₁, L₂ and L₃ or slight differences among referencepotentials V₁, V₂ and V₃ caused thereby. If this phenomenon occurs, thenexpected characteristics are not obtained, besides there is a dangerthat an element might be destroyed by the concentrated currents. On theother hand, in the switching power supply apparatus of the currentfeedback control mode, it is comparatively easy to implement themulti-phase operation in which a plurality of power supply circuits areconnected in parallel and caused to operate. This is because it ispossible to uniformly distribute the currents I₁, I₂ and I₃, which flowthrough the inductors by current feedback control and prevent currentconcentration. However, there has been no example in which multi-phaseoptimal transitional response operation is implemented by the currentfeedback control until now. This is because the calculation error of thepeak current preset value is so large and time consuming as to beunallowable in the optimal transitional response operation.

In the present embodiment, multi-phase implementation inclusive of thetransitional response operation is implemented by individually detectingand storing the inductor currents and averaging them. According to thepresent embodiment, therefore, a large supply current can be obtainedwith small-sized elements, especially small inductors by connecting aplurality of DC power supplies in parallel. Therefore, it is possible toreduce the size and cost of the power supply apparatus.

While certain embodiments have been described, these embodiments havebeen presented by way of example only, and are not intended to limit thescope of the inventions. Indeed, the novel methods and systems describedherein may be embodied in a variety of other forms; furthermore, variousomissions, substitutions and changes in the form of the methods andsystems described herein may be made without departing from the spiritof the inventions. The accompanying claims and their equivalents areintended to cover such forms or modifications as would fall within thescope and spirit of the inventions.

1. A switching power supply apparatus comprising: a power circuit unit,the power circuit unit comprising a first switching element connected ata first end to a first end of a power supply, an inductor connected at afirst end thereof to a second end of the first switching element, afirst output terminal connected to a second end of the inductor, asecond output terminal connected to a second end of the power supply, acapacitor connected between the first output terminal and the secondoutput terminal, and a second switching element connected between thesecond end of the power supply and a second end of the first switchingelement; and an optimal response multi-mode digital current program modecontrol unit, the optimal response multi-mode digital current programmode control unit comprising an error signal generator which generatesan error signal according to a potential difference between an outputvoltage and a preset voltage, an inductor current detector which detectsand amplifies an inductor current, a mode detector which detects anabrupt change of the output voltage and switches an operation mode; apeak/valley detector & optimal current difference calculator whichdetects an extreme value of the output voltage and calculates a chargequantity lost by the capacitor; and an inductor current detectionstorage circuit which detects and stores an inductor current obtainedwhen an extreme value of the output voltage is detected, and on-offcontrol of the first switching element or the second switching elementbeing exercised according to control of the mode detector, thepeak/valley detector & optimal current difference calculator, and theinductor current detection storage circuit.
 2. The switching powersupply apparatus according to claim 1, wherein the power circuit unit isprovided in plural sets, and wherein the inductor current detectionstorage circuit and the on-off control share the single optimal responsemulti-node digital current program mode control unit.
 3. The switchingpower supply apparatus according to claim 1, further comprising aprogrammable current protection and limiter which detects that theinductor current has exceeded a preset value and which switches theon-off state of the first switching element and the second switchingelement temporarily.
 4. The switching power supply apparatus accordingto claim 1, wherein the extreme value is a minimum or maximum value. 5.The switching power supply apparatus according to claim 1, wherein thefirst switching element and the second switching element are MOStransistors.
 6. A switching power supply control apparatus whichcontrols a first switching element connected at a first terminal thereofto a high potential terminal of an external input power supply and asecond switching element connected at a first terminal thereof to a lowpotential terminal of the external input power supply and connected at asecond terminal thereof to a second terminal of the first switchingelement, the switching power supply apparatus comprising an optimalresponse multi-mode digital current program mode control unit, theoptimal response multi-mode digital current program mode control unitcomprising an error signal generator which generates an error signalaccording to a potential difference between an output voltage and apreset voltage, and an inductor current detector which detects andamplifies an inductor current, wherein when a potential differencebetween the output voltage and the preset voltage has exceeded apredetermined voltage value, the first switching element is turned on,an optimal peak current value is calculated from an inductor currentvalue at timing of the output voltage assuming an extreme value, andwhen the inductor current has reached the optimal peak current value,the first switching element is turned off.
 7. The switching power supplycontrol apparatus according to claim 6, wherein the first switchingelement, the second switching element and inductor current detector areprovided in plural sets, and wherein the plurality of the inductorcurrent detectors share the single optimal response multi-node digitalcurrent program mode control unit.
 8. The switching power supply controlapparatus according to claim 6, wherein the extreme value is a minimumof maximum value, depending on the type of load transition.
 9. Aswitching power supply control apparatus which controls a firstswitching element connected at a first terminal thereof to a highpotential terminal of an external input power supply and a secondswitching element connected at a first terminal thereof to a lowpotential terminal of the external input power supply and connected at asecond terminal thereof to a second terminal of the first switchingelement, the switching power supply apparatus comprising an optimalresponse multi-mode digital current program mode control unit, theoptimal response multi-mode digital current program mode control unitcomprising an error signal generator which generates an error signalaccording to a potential difference between an output voltage and apreset voltage, and an inductor current detector which detects andamplifies an inductor current, wherein when a potential differencebetween the output voltage and the preset voltage has exceeded apredetermined voltage value, the first switching element is turned on,an optimal peak current value is calculated from an inductor currentvalue at timing of the output voltage assuming an extreme value and thepotential difference between the extreme value of the output voltage andthe preset voltage, and when the inductor current has reached theoptimal peak current value, the first switching element is turned off.10. The switching power supply control apparatus according to claim 9,wherein the first switching element, the second switching element andinductor current detector are provided in plural sets, and wherein theplurality of the inductor current detectors share the single optimalresponse multi-node digital current program mode control unit.
 11. Theswitching power supply control apparatus according to claim 9, whereinthe extreme value is a minimum or maximum value, depending on the typeof the load transient.
 12. A switching power supply control apparatuswhich controls a first switching element connected at a first terminalthereof to a high potential terminal of an external input power supplyand a second switching element connected at a first terminal thereof toa low potential terminal of the external input power supply andconnected at a second terminal thereof to a second terminal of the firstswitching element, and which has a current program mode for steady loadstate and an optimum load response mode, the switching power supplyapparatus comprising an optimal response multi-mode digital currentprogram mode control unit, the optimal response multi-mode digitalcurrent program mode control unit comprising an error signal generatorwhich generates an error signal according to a potential differencebetween an output voltage and a preset voltage, and an inductor currentdetector which detects and amplifies an inductor current, wherein when apotential difference between the output voltage and the preset voltagehas exceeded a predetermined voltage value, a transition from thecurrent program mode to the optimal load response mode is conducted andthe first switching element is turned on, an inductor current value attiming of the output voltage assuming an extreme value is retained as asteady peak current value, at the same time, an optimal peak currentvalue is calculated from the inductor current value and the potentialdifference between the extreme value of the output voltage and thepreset voltage, when the inductor current has reached the optimal peakcurrent value, the first switching element is turned off, and at timingof the output voltage returning to the preset voltage, transition to thecurrent program mode having the steady peak current value as a presetcurrent is conducted.
 13. The switching power supply control apparatusaccording to claim 12, which controls plural sets of the first switchingelement, then second switching element, wherein the plural inductorcurrent detectors share the single optimal response multi-node digitalcurrent program mode control unit.
 14. The switching power supplycontrol apparatus according to claim 12, wherein the extreme value is aminimum value.
 15. A switching power supply apparatus comprising anoptimal response multi-mode digital current program mode control unit,the optimal response multi-mode digital current program mode controlunit comprising an output potential monitor and an inductor currentdetection storage circuit, the output potential monitor detecting timingof an output potential assuming an extreme value, and upon receiving asignal of the detected timing, the inductor current detection storagecircuit detecting an inductor current and stores a value of the inductorcurrent.
 16. The switching power supply control apparatus according toclaim 15, wherein the inductor current detector is provided in pluralsets, and wherein the plurality of the inductor current detectors sharethe single optimal response multi-node digital current program modecontrol unit.
 17. The switching power supply apparatus according toclaim 15, wherein the extreme value is a minimum or maximum value,depending on the type of the load transient.
 18. A switching powersupply apparatus comprising: a power circuit unit, the power circuitunit comprising a first switching element connected at a first end to afirst end of a power supply, an inductor connected at a first endthereof to a second end of the first switching element, a first outputterminal connected to a second end of the inductor, a second outputterminal connected to a second end of the power supply, a capacitorconnected between the first output terminal and the second outputterminal, and a second switching element connected between the secondend of the power supply and a second end of the first switching element;and an optimal response multi-mode digital current program mode controlunit, wherein the optimal response multi-mode digital current programmode control unit comprises: a peak/valley detector & optimal currentdifference calculator which detects a potential difference between thefirst output terminal and the second output terminal, detects that thepotential difference has reached an extreme value, generates apeak/valley detection signal, and calculates an optimal peak currentdifference; a mode detector which detects that an error signalindicating a difference between a potential difference obtained betweenthe first output terminal and the second output terminal and a presetvoltage is greater than a certain preset value and generates a modeswitching signal; an error multiplier which outputs a signal obtained byamplifying the error signal; a first mode switch which selects either anoutput of the error multiplier or an output of the peak/valley detector& optimal current difference calculator according to the mode switchingsignal, and outputs the selected result as a latest optimal peak currentdifference signal; a successive approximation DA converter which addsthe optimal peak current difference signal and the optimal peak currentsignal of last time together, and thereby calculates the latest optimalpeak current signal; an inductor current detector which obtains aninductor current signal by amplifying the inductor current; a sample andhold circuit which takes in and retains the inductor current signalaccording to the peak/valley detection signal; a second mode switchwhich selects and retains either the inductor current signal output bythe sample and hold circuit or an output signal of the successiveapproximation DA converter according to the peak/valley detectionsignal, and outputs the latest optimal peak current value; an optimalcurrent value retaining capacitor which stores the latest optimal peakcurrent value to make it possible for the successive approximation DAconverter to find an optimal peak current value of last time in nextcycle operation; a differential amplifier which is supplied at anon-inverting input terminal thereof with the inductor current signaland supplied at an inverting input terminal thereof with the latestoptimal peak current value, which comprises the inductor current signalwith the latest optimal peak current value, amplifies a differencebetween the inductor current signal and the latest optimal peak currentvalue, and generates a reset signal; an SR latch which is supplied at areset terminal thereof with the reset signal and supplied at a setterminal thereof with a clock signal, and which generates a switchingelement control signal; and a drive circuit which is supplied with theswitching element control signal and which exercises on/off control onthe first switching element and the second switching element.
 19. Theswitching power supply apparatus according to claim 18, wherein thepower circuit unit is provided in plural sets, and wherein thesuccessive approximation DA converter, the inductor current detector,the sample and hold circuit, the second mode switch, the optimal currentvalue retaining capacitor, the differential amplifier, the SR latch andthe drive circuit share the single optimal response multi-node digitalcurrent program mode control unit.
 20. The switching power supplyapparatus according to claim 18, wherein the extreme value is a minimumor maximum value, depending on the type of the load transient.
 21. Theswitching power supply apparatus according to claim 18, wherein thefirst switching element and the second switching element are MOStransistors.